Apparatus and method for sensing of isolated output

ABSTRACT

A power converter includes a current controller coupled to an energy transfer element to selectively enable a first, second or third current in the current controller. The first current is substantially zero, the second current is greater than the third current, and the third current is greater than the first current. The third current only partially discharges a capacitance coupled to the energy transfer element and the current controller. A control circuit is to be coupled to the current controller to selectively enable the first, second or third current in the current controller. A first feedback circuit is coupled to generate a first feedback signal while the first current is enabled by the current controller after a full discharge pulse. A second feedback circuit is coupled to generate a second feedback signal while the first current is enabled in the controller after a partial discharge pulse.

REFERENCE TO PRIOR APPLICATION(S)

This is a continuation of U.S. application Ser. No. 13/398,000, filedFeb. 16, 2012, now pending, which is a continuation of U.S. applicationSer. No. 12/770,478, filed Apr. 29, 2010, now U.S. Pat. No. 8,144,487.U.S. application Ser. Nos. 13/398,000 and 12/770,478 are herebyincorporated by reference.

BACKGROUND INFORMATION

1. Field of the Disclosure

This invention is related to the control of switched-mode powersupplies. Specifically, it is related to low-cost power supplies withregulated isolated outputs that must meet standards for maximum powerconsumption when the output has no load, and yet must keep the outputwithin specified limits when a load is suddenly applied.

2. Background

Low-cost solutions to regulate an isolated output voltage of a switchingpower supply typically rely on the magnetic coupling between isolatedwindings of an energy transfer element to provide information about theoutput to a control circuit. The control circuit typically receives asignal representative of the output voltage immediately after aswitching event that delivers energy to the output. The signal istypically received from a primary-referenced winding of an energytransfer element instead of from an optocoupler. This type of control isoften referred to as “primary-side control” or control usingprimary-side feedback.

Although these solutions eliminate the cost and the power consumed by anoptocoupler, they cannot sense the output voltage in the absence ofswitching. A problem arises when the load on the output of the powersupply approaches zero. The power supply must provide the specifiedregulated output voltage but almost no power. Under such conditions, thepower lost in the operation of the power supply itself is a significantpart of the total power consumed. Requirements to limit the consumptionof power by the power supply under conditions of near zero externalloading discourage the use of a dummy internal load in a power supply. Adummy internal load, sometimes called a pre-load, can be useful in apower supply to help provide overvoltage protection, improve regulationbetween multiple outputs, and prevent the switching frequency from goingbelow a minimum value. In particular, a dummy internal load is a smallpermanent minimum load inside a power supply. However, a penalty forusing a dummy internal load is that the power supply becomes lessefficient because the dummy load dissipates power that is not measuredas output power. Also, the controller has to switch more often to powerthe additional internal load, which results in the power supplyconsuming additional power even though there is no load connected to thepower supply's output. To avoid these drawbacks, it is useful toincrease the time between switching events under no-load conditions toreduce the losses inherent in switching. However, under such conditions,the controller is unable to sense the output voltage during therelatively long intervals between switching events.

When a substantial load is suddenly applied to the output during one ofthe relatively long intervals between switching events, the outputvoltage can easily fall outside the specified limits of regulationbefore the controller is able to respond to the condition. A typicalremedy for such a condition is the addition of costly bulk capacitanceto the output to provide the energy required by a load that could beapplied during the time when the controller cannot sense the output.

BRIEF DESCRIPTION OF THE DRAWINGS

Non-limiting and non-exhaustive embodiments of the present invention aredescribed with reference to the following figures, wherein likereference numerals refer to like parts throughout the various viewsunless otherwise specified.

FIG. 1 shows an example power converter including a controller inaccordance with the teachings of the present invention that providessensing of an isolated output.

FIG. 2 is an example of a power converter including a controller thatuses a winding on a coupled inductor to sense output voltage inaccordance with the teachings of the present invention and that providessensing of an isolated output.

FIG. 3 shows voltage and current waveforms from an example powerconverter that illustrates the operation of a controller in accordancewith the teachings of the present invention that provides for sensing ofan isolated output voltage.

FIG. 4 shows one example of a power converter that illustrates oneexample of a current controller in accordance with the teachings of thepresent invention.

FIG. 5 shows one example of a power converter that illustrates anotherexample of a current controller, which uses a transistor for sensing ofan isolated output voltage in accordance with the teachings of thepresent invention.

FIG. 6 is flow diagram that shows one example method to control a powerconverter in accordance with the teachings of the present invention thatprovides for sensing of an isolated output voltage.

DETAILED DESCRIPTION

Methods and apparatuses for implementing a power supply controller thatprovide relatively low cost solutions that accomplish sensing of anisolated output of a power converter are disclosed. In the followingdescription, numerous specific details are set forth in order to providea thorough understanding of the present invention. It will be apparent,however, to one having ordinary skill in the art that the specificdetail need not be employed to practice the present invention. In otherinstances, well-known materials or methods have not been described indetail in order to avoid obscuring the present invention.

Reference throughout this specification to “one embodiment”, “anembodiment”, “one example” or “an example” means that a particularfeature, structure or characteristic described in connection with theembodiment or example is included in at least one embodiment of thepresent invention. Thus, appearances of the phrases “in one embodiment”,“in an embodiment”, “one example” or “an example” in various placesthroughout this specification are not necessarily all referring to thesame embodiment or example. Furthermore, the particular features,structures or characteristics may be combined in any suitablecombinations and/or subcombinations in one or more embodiments orexamples. Particular features, structures or characteristics may beincluded in an integrated circuit, an electronic circuit, acombinational logic circuit, or other suitable components that providethe described functionality. In addition, it is appreciated that thefigures provided herewith are for explanation purposes to personsordinarily skilled in the art and that the drawings are not necessarilydrawn to scale.

FIG. 1 is a schematic diagram that shows generally one example of aswitching power converter 100 that uses a flyback topology in accordancewith the teachings of the present invention. In the illustrated example,power converter 100 is shown as a power supply having flyback topologyfor explanation purposes. It is noted, however, that there are manyother known topologies and configurations for switching power supplies.It is appreciated that the example flyback topology illustrated in FIG.1 is adequate for explaining principles in accordance with the teachingsof the present invention and that the principles may apply also to othertypes of switching regulators in accordance with the teachings of thepresent invention. Details that will be discussed in greater detailbelow are omitted from FIG. 1 to avoid obscuring teachings in accordancewith the present invention.

The example power converter in FIG. 1 controls the transfer of energyfrom an unregulated input voltage V_(IN) 102 at the input of the powerconverter 100 to a load 122 at the output of the power converter 100.The input voltage V_(IN) 102 is coupled to an energy transfer element T1105 and a current controller 148. In the example of FIG. 1, the energytransfer element T1 105 is a coupled inductor, sometimes referred to asa transformer, with a primary winding 108 and a secondary winding 112.In the example of FIG. 1, primary winding 108 may be considered an inputwinding, and secondary winding 112 may be considered an output winding.A clamp circuit 104 is coupled to the primary winding 108 of the energytransfer element T1 105 to control the maximum voltage on the currentcontroller 148.

In the example of FIG. 1, input voltage V_(IN) 102 is positive withrespect to an input return 110, and output voltage V_(O) 120 is positivewith respect to an output return 124. The example of FIG. 1 showsgalvanic isolation between the input return 110 and the output return124 because input return 110 and output return 124 are designated bydifferent symbols. In other words, a dc voltage applied between inputreturn 110 and output return 124 will produce substantially zerocurrent. Therefore, circuits electrically coupled to the primary winding108 are galvanically isolated from circuits electrically coupled to thesecondary winding 112.

In the illustrated example, current controller 148 either conductscurrent or does not conduct current in response to a control circuit 144that is included in a controller 142. Current controller 148 andcontroller 142 may include integrated circuits and discrete electricalcomponents. In some examples, current controller 148 and controller 142may be integrated together in a single monolithic integrated circuit.

In the example of FIG. 1, current controller 148 controls a currentI_(P) 126 in response to controller 142 to meet a specified performanceof the power converter 100. In operation, current controller 148produces pulsating current in primary winding 108 and in secondarywinding 112. Current in secondary winding 112 is rectified by rectifierD 114 and then filtered by capacitor C1 116 to produce a substantiallyconstant output voltage V_(O) 120 or output current I_(O) 118 at theload 122. The operation of current controller 148 also produces a timevarying voltage V_(P) 106 between the ends of primary winding 108. Bytransformer action, a scaled replica of the voltage V_(P) is producedbetween the ends of secondary winding 112, the scale factor being theratio that is the number of turns of secondary winding 112 divided bythe number of turns of primary winding 108.

The example illustrated in FIG. 1 shows a capacitor C_(P) 150 in brokenlines at the node between one end of primary winding 108 and currentcontroller 148. Capacitor C_(P) 150 in the example of FIG. 1 representsall the capacitance that couples to current controller 148. CapacitorC_(P) 150, which could be referred to as the primary switching nodecapacitance, may include natural capacitance internal to energy transferelement T1 105 as well as the natural internal capacitance of currentcontroller 148. Capacitor C_(P) 150 may also include discrete capacitorsplaced intentionally in various parts of the circuit to filter noise andto slow transitions of switching voltages. Capacitor C_(P) 150 has avoltage V_(CP) 128 that is the voltage at one end of primary winding 108with respect to the input return 110. The importance of capacitor C_(P)150 will become apparent later in this disclosure.

In the example of FIG. 1, a sensor 134 receives a sense signal 132 thatis representative of the output quantity to be regulated at the outputof power converter 100. The output quantity to be regulated bycontroller 142 is typically the output voltage V_(O) 120, but in someexamples is the output current I_(O) 118, and in other examples may be acombination of output voltage V_(O) 120 and output current I_(O) 118.Controller 142 receives a feedback signal U_(FB) 136 from sensor 134.Feedback signal U_(FB) 136 may be either a voltage or a current.

Since circuits electrically coupled to the secondary winding 112 aregalvanically isolated from the circuits electrically coupled to theprimary winding 108, either the sense signal 132 is galvanicallyisolated from the load 122, or sensor 134 provides galvanic isolationbetween sense signal 132 and controller 142. In other words, galvanicisolation may reside in either the sensor 134 or in another part of thepath of the sense signal 132 not shown in FIG. 1.

In the example of FIG. 1, controller 142 receives a current sense signal130 that is representative of the current I_(P) 126. Current sensesignal 130 may be either a voltage or a current and may be obtainedusing known methods. For example, current sense signal 130 may be theoutput of a current transformer, the voltage across a current senseresistor, or the voltage across the on-resistance of a metal oxidefield-effect transistor MOSFET that conducts either the entire currentI_(P) 126 or a portion of the current I_(P) 126.

In the example of FIG. 1, controller 142 receives feedback signal U_(FB)136 and current sense signal 130 to produce a mode select signal 146that is received by current controller 148. In one example, currentcontroller 148 may have three modes of operation. A first mode may beone that does not conduct current, such that current I_(P) 126 issubstantially zero when current controller 148 is the first mode. Asecond mode may be one that conducts as much current as externalcircuitry allows, such as for example the condition where the current inthe primary winding 108 of energy transfer element T1 105 is determinedby the input voltage V_(IN), the inductance of primary winding 108, andthe time that current controller 148 remains in the second mode. A thirdmode may be one that restricts conduction of current to a relativelysmall value during the time the current controller 148 remains in thethird mode in accordance with the teachings of the present invention. Inone example, the relatively small value for the current is a constantcurrent value that is substantially less than the current value duringthe second mode. In one example, the relatively small constant currentvalue of the third mode is 5 percent of the maximum current conducted inthe second mode.

In the example of FIG. 1, feedback signal U_(FB) 136 has substantiallydifferent characteristics that depend on the changes in modes of currentcontroller 148 in accordance with the teachings of the presentinvention. For example, when current controller 148 changes between thesecond mode and the first mode, the feedback signal U_(FB) 136 containsfeatures that are not present when current controller 148 changesbetween the third mode and the first mode. Therefore, controller 142includes a first feedback circuit 138 and a second feedback circuit 140to interpret the feedback signal U_(FB) 136 appropriately for thedifferent modes of current controller 148 in accordance with theteachings of the present invention. Examples of other controllers mayinclude more than two feedback circuits as required to interpretfeedback signals that arise from different modes of operation.

In the example of FIG. 1, control circuit 144 included in controller 142receives signals from first feedback circuit 138 and second feedbackcircuit 140 to control the output of the power converter as desired.Feedback circuits included in controller 142 may use any analog anddigital circuits such as filter circuits, sample and hold circuits, andcomparators, to extract necessary information from feedback signalU_(FB) 136. Control circuit 144 included in controller 142 may use anyanalog and digital circuits, such as oscillators, comparators, digitallogic, and state machines, or the like, to interpret and respond asrequired to information received from the feedback circuits.

FIG. 1 shows mode select signal 146 as a single line that in anotherexample may represent several individual analog or digital signals. Forexample, two binary digital signals lines for control signal 146 mayselect as many as four distinct modes of current controller 148 inaccordance with the teachings of the present invention.

FIG. 2 is a schematic diagram that shows another example of a switchingpower converter 200 that uses a flyback topology in accordance withteachings of the present invention. The example of FIG. 2 includes acoupled inductor 205 that has a primary winding 108, a secondary winding112, and a bias winding 210. Bias winding 210 may also be referred to asan auxiliary winding. In one example, bias winding 210 in FIG. 2 is thesensor 134 introduced in FIG. 1 that produces feedback signal U_(FB)136. Bias winding 210 produces a voltage V_(B) 215 that is responsive tothe output voltage V_(O) 120 when rectifier D1 114 on secondary winding112 conducts. Sense signal 132, shown in FIG. 1 but not visible in FIG.2, is the magnetic flux that couples bias winding 210 to secondarywinding 112 of the coupled inductor 205. In another example, biaswinding 210 may also provide a source of power to the circuits withincontroller 142.

It is appreciated that many variations are possible in the use of a biaswinding to sense an output voltage and for providing sensing while alsoproviding power to a controller with galvanic isolation. For example, abias winding may apply a rectifier and a capacitor similar to rectifierD1 114 and capacitor C1 116, respectively, to produce a dc bias voltagewhile providing an ac feedback signal from the anode of the rectifier.As such, additional passive components such as resistors may be used onthe bias winding to scale the voltage from the winding to a value thatis more suitable to be received by controller 142.

Use of bias winding 210 to sense output voltage V_(O) 120 has theadvantages of providing galvanic isolation between output voltage V_(O)120 and controller 142 without the expense of an optocoupler. Adisadvantage of using a winding on coupled inductor 205 to sense outputvoltage V_(O) 120 is that the voltage V_(B) 215 at bias winding 210 isrepresentative of output V_(O) 120 only when output rectifier D1 114 isconducting, whereas an optocoupler can provide continuous sensing ofoutput voltage V_(O) 120. Output rectifier D1 114 is conducting onlywhile there is a pulse of current in secondary winding 112. Therefore,the time between pulses of current in secondary winding 112 is the timewhen controller 142 cannot sense output voltage V_(O) 120. In otherwords, in contrast to sensing output voltage V_(O) 120 continuously withan optocoupler, sensing output voltage V_(O) 120 with a winding oncoupled inductor 205 is limited to pulses that may not occur oftenenough to provide the necessary information for the desired control ofoutput voltage V_(O) 120. Since secondary winding 112 has a pulse ofcurrent only after primary winding 108 has a pulse of current, it isdesirable to decrease the time between pulses of current in primarywinding 108 so that controller 142 can sense output voltage V_(O) 120more often.

The rate and magnitude of pulsating current in primary winding 108 iscontrolled by controller 142 to provide the power required to maintainthe desired output voltage V_(O) 120 over a range of values of load 122.As the load approaches zero, less current in primary winding 108 isneeded to maintain the desired output voltage V_(O) 120. As such,controllers may reduce the magnitude of the current in primary winding108 as well as increase the time between pulses of current.

Controller 142 may produce pulses of current in primary winding 108 byproviding current controller 148 with a mode select signal 146 thatchanges current controller 148 from the first mode to the second mode,allowing current I_(P) 126 to increase with a linear slope to a desiredmaximum before returning to the first mode. Operation of currentcontroller 148 in the second mode fully discharges capacitor C_(P) 150so that voltage V_(P) 106 on primary winding 108 is equal to inputvoltage V_(IN) 102.

All the energy stored on capacitor C_(P) 150 is lost when currentcontroller 148 operates in the second mode, even if the maximum currentI_(P) 126 is allowed to increase to the lowest practical value beforereturning to the first mode. The only way to reduce the power dissipatedfrom the full discharge of capacitor C_(P) 150 is to increase the timebetween discharges. In other words, increasing the time between pulsesof primary current will reduce the power lost in the power converter asthe load approaches zero at the expense of increasing the time where thecontroller 142 cannot sense the output voltage V_(O) 120. As aconsequence, a sudden increase in the load 122 may reduce the outputvoltage V_(O) 120 to an unacceptable value before controller 142 cansense the voltage and respond to it.

A solution is discussed below that allows the controller 142 to sensethe output voltage V_(O) 120 frequently enough to respond adequately toa sudden increase in the load 122 while also reducing power dissipationat near zero load. This solution produces pulses of current in primarywinding 108 without fully discharging capacitor C_(P) 150. The solutionis realized by the introduction of a third mode of operation for currentcontroller 148 in accordance with the teachings of the presentinvention. In one example, the third mode of current controller 148operates to put only enough current into primary winding 108 so thatoutput rectifier D1 114 will conduct after current controller 148returns to the first mode. The third mode of current controller 148conducts current with a sufficiently low magnitude and duration to putthe desired current into the primary winding while only partiallydischarging capacitor C_(P) 150 in accordance with the teachings of thepresent invention. The determination of the proper value of current fora given application is discussed in detail later in this disclosure.

FIG. 3 shows voltage and current waveforms from the example powerconverter of FIG. 2 that illustrates the operation of an examplecontroller in accordance with the teachings of the present inventionthat provides sensing of an isolated output voltage. As shown in thedepicted example, pulses of current I_(P) 126 that fully dischargecapacitor C_(P) 150 begin at times t₀ 310, t_((N+1)) 350, and t_((N+2))360. Pulses of current I_(P) 126 that partially discharge capacitorC_(P) 150 begin at times t₁ 320, t₂ 330, and t_(N) 340 in accordancewith the teachings of the present invention.

The distinction between pulses of current I_(P) 126 that fully dischargecapacitor C_(P) 150 and pulses of current I_(P) 126 that partiallydischarge capacitor C_(P) 150 is clear in the waveform of V_(CP) 128that is the voltage on capacitor C_(P) 150. Capacitor C_(P) 150 is fullydischarged when the voltage V_(CP) 128 is substantially zero. CapacitorC_(P) 150 is only partially discharged when the voltage V_(CP) 128remains substantially greater than zero when current I_(P) 126 isgreater than zero.

As shown, at the end of each full-discharge pulse and eachpartial-discharge pulse of current I_(P) 126, voltage V_(CP) 128 risesabove the input voltage V_(IN) 102 while energy from the energy transferelement (e.g., energy transfer element T1 105 in FIG. 1 and coupledinductor 205 in FIG. 2) charges capacitor C_(P) 150. Voltage V_(CP) 128rises until the output rectifier D1 114 conducts, clamping the voltageV_(CP) to the input voltage V_(IN) plus the reflected output voltageV_(OR), where the reflected output voltage V_(OR) is the voltage onsecondary winding 112 multiplied by the number of turns on primarywinding 108 and divided by the number of turns on the secondary winding112.

As shown, voltage V_(CP) 128 remains clamped at the value of V_(IN) plusV_(OR) until output rectifier D1 114 stops conducting, which happenswhen the current from secondary winding 112 falls to zero. The energystored in capacitor C_(P) 150 that raised its voltage above V_(IN) 102then dissipates in a decaying oscillation with the self-inductance ofprimary winding 108 and the effective parasitic resistance of thecircuit.

FIG. 3 also shows the voltage V_(B) 215 in FIG. 2 that provides feedbacksignal U_(FB) 136 to controller 142. Controller 142 may sense inputvoltage V_(IN) 102 as well as output voltage V_(O) 120 from bias voltageV_(B) 215. During full-discharge pulses of current I_(P) 126, biasvoltage V_(B) 215 goes negative to a magnitude V_(INS) that isrepresentative of input voltage V_(IN) 102. After a full-discharge pulseof current I_(P) 126, output rectifier D1 114 conducts to allow sensingof output voltage V_(O) 120 as a positive voltage V_(OS) on bias winding210 that is representative of output voltage V_(O) 120. After a pulse ofcurrent I_(P) 126 that only partially discharges capacitor C_(P) 150,output rectifier D1 114 conducts just enough to allow sensing of outputvoltage V_(O) 120 with a decaying oscillation of bias voltage V_(B) 215as shown in FIG. 3.

In one example, when the load 122 is large enough to requirefull-discharge pulses of current I_(P) 126 to maintain output voltageV_(O) 120 at a desired value, the full-discharge pulses may occur asoften as every switching period T_(S). An example switching period T_(S)is the time between t_((N+1)) 350 and t_((N+1)) 360 in FIG. 3.Typically, light to moderate loads may require patterns offull-discharge pulses separated by several switching periods of nocurrent pulses.

In the example, when the load 122 is near zero, only partial-dischargepulses are used to sense the output voltage V_(O) 120 at intervals muchshorter than the period between full-discharge switching pulses inaccordance with the teachings of the present invention. It is notnecessary to use partial-discharge pulses to sense the output voltageV_(O) 120 when the load 122 is sufficiently greater than zero becausefull-discharge pulses occur often enough at loads sufficiently greaterthan zero to provide adequate sensing of the output voltage. Thepartial-discharge pulses may be considered wake-up pulses that determinewhether or not a full-discharge pulse is required. The times betweenpartial-discharge pulses may be considered wake-up periods.

In the illustrated example, an example wake-up period T_(W1) is the timebetween t₁ 320 and t₂ 330 in FIG. 3. In one example, wake-up periodT_(W1) is 16 switching periods T_(S). In another example, wake-up pulsesmay be separated by wake-up periods of different durations. The firstpartial-discharge pulse in a train of partial-discharge pulses mayfollow a full-discharge pulse by a period that is different from anywake-up period within a train of wake-up pulses. FIG. 3 shows a periodT₁ that is the time between t_(O) 310 at the start of a full-dischargepulse and time t₁ 320 that is the start of the first partial-dischargepulse in a train of partial-discharge pulses. In one example, the periodT₁ is 9 switching periods whereas T_(W1) is 16 switching periods.

In one example, control circuit 144 included in controller 142 maydetermine the need for full-discharge pulses, partial discharge pulses,or no pulse within a switching period T_(S) according to the value offeedback signal U_(FB) 136 immediately after each pulse of current I_(P)126 in accordance with the teachings of the present invention. Forexample, if a sequence of full-discharge pulses causes the sensed outputvoltage V_(OS) to rise beyond a first threshold value, control circuit144 may set mode select signal 146 such that current controller 148conducts no current for several subsequent switching periods T_(S). Ifsensed output voltage V_(OS) remains above the first threshold valueafter the next full-discharge pulse, controller 142 may conclude thatthe load is near zero and begin using partial discharge pulses to sensethe output voltage V_(O) 120. The example of FIG. 3 illustrates anexample in which control circuit 144 determined that sensed outputvoltage V_(OS) after the partial-discharge pulse at time T_(N) 340 wastoo low, requiring consecutive full-discharge pulses at times t_((N+1))350 and t_((N+1)) 360.

In the example of FIG. 2, control circuit 144 interprets a signalreceived from first feedback circuit 138 after a full-discharge pulse,and control circuit 144 interprets a signal received from secondfeedback circuit 140 after a partial-discharge pulse in accordance withthe teachings of the present invention. In the example of FIG. 2, firstfeedback circuit 138 samples feedback signal U_(FB) 136 during the timewhen output rectifier D1 114 is conducting. In the example of FIG. 2,second feedback circuit 138 samples feedback signal U_(FB) 136 during adecaying oscillation of the bias voltage V_(B) 215 after outputrectifier D1 114 stops conducting.

In the example, the peaks of the decaying oscillation in the biasvoltage V_(B) 215 are representative of output voltage V_(O) 120 afteroutput rectifier D1 114 stops conducting because capacitor C_(P) 150charges to a value representative of output voltage V_(O) 120 whenoutput rectifier D1 114 conducts after a partial-discharge pulse. Themaximum voltage on capacitor C_(P) 150 sets the initial condition forthe decaying oscillation after output rectifier D1 114 stops conducting.Therefore, each peak in the decaying oscillation of bias voltage V_(B)215 is determined by the maximum voltage on capacitor C_(P) 150 after apartial-discharge pulse.

In the example of FIG. 2, first feedback circuit 138 samples feedbacksignal U_(FB) 136 to regulate output voltage V_(O) 120 over a wide rangeof loads. In contrast to first feedback circuit 138, second feedbackcircuit 140 in one example does not sample feedback signal U_(FB) 136 toregulate output voltage V_(O) 120 over a wide range of load. Instead,second feedback circuit 140 in the example is used only to determinewhether or not there has been a sufficient change in the output voltageV_(O) 120 during a train of partial-discharge pulses to require a changein operating mode in accordance with the teachings of the presentinvention.

Specifically, in one example, second feedback circuit 140 holds thevalue of the second peak in the decaying oscillation of feedback signalU_(FB) 136, as illustrated for example in FIG. 3 with the decayingoscillations in V_(B) 215 after the first partial-discharge pulse in atrain of consecutive partial-discharge pulses, and compares it tosamples of the second peak in the decaying oscillation of feedbacksignal U_(FB) 136 after each subsequent partial-discharge pulse in thetrain of consecutive partial-discharge pulses. When the value of asubsequent sample is less than the value of the first sample by athreshold value, control circuit 144 determines that output voltageV_(O) 120 is too low, and sets mode select signal 146 to start asequence of full-discharge pulses. It is appreciated that any peak valuein the decaying oscillation may be sampled for use in the comparison. Inone example, the second peak value may be a preferred peak because ithas the highest magnitude and is relatively free from noise anddistortion that may be present on the first peak while the outputrectifier D1 114 is conducting. In one example, the threshold value is30 millivolts.

In one example, the magnitude and duration of a partial-discharge pulseare just sufficient to allow output rectifier D1 114 to conduct at theend of the partial-discharge pulse. In another example, the magnitudeand duration of a partial-discharge pulse are more than sufficient toallow output rectifier D1 144 to conduct at the end of thepartial-discharge pulse. The output voltage V_(O) 120 may be sensed withgreater accuracy when output rectifier D1 144 is allowed to conductuntil a transient voltage associated with non-ideal coupling of thewindings of coupled inductor 205 reduces to a negligible value. Thenon-ideal coupling, sometimes quantified as a leakage inductance, mayproduce a voltage between output rectifier D1 144 and secondary winding112 when diode D1 144 begins to conduct. The transient voltage owing toleakage inductance may also distort the first peak of the decayingoscillation. Therefore, it is desirable to allow the voltage from theleakage inductance to reduce to a negligible value so that capacitorC_(P) 150 charges to a voltage that more accurately represents outputvoltage V_(O) 120 before output rectifier D1 144 stops conducting. It isalso desirable not to sample the first peak of the decaying oscillationto avoid distortion from the effects of leakage inductance.

In one example, the magnitude of the partial-discharge pulse is 16milliamperes whereas the peak current of a full-discharge pulse is 250milliamperes. As such, the energy transferred to the output by thepartial-discharge pulse may be considered insignificant in comparison tothe energy transferred to the output by the full-discharge pulse becausethe energy transferred is proportional to the square of the peak currentin primary winding 108. It will be appreciated that since thepartial-discharge pulse may transfer finite energy to the output,controllers that have a minimum switching frequency, however small, mayrequire the power supply to have a dummy internal load to keep theoutput voltage V_(O) 120 from going higher than desired as the outputcurrent I_(O) 118 approaches zero.

It may be determined either analytically or experimentally that there isa magnitude and a duration for a partial-discharge pulse that gives aminimum power loss in the power converter for a particular set ofcircumstances. The duration of the partial-discharge pulse is typicallyless than half of one period of the decaying oscillation of feedbacksignal U_(FB) 136 as illustrated in FIG. 3. In one example, the durationof the partial-discharge pulse is approximately one quarter of oneperiod of the decaying oscillation of feedback signal U_(FB) 136. In oneexample where the inductance of the primary winding 108 of a coupledinductor is 2.2 millihenries, capacitor C_(P) 150 is approximately 70picofarads, the partial discharge pulse is 16 milliamperes for aduration of approximately 600 nanoseconds. It will be appreciated thatin one example control circuit 144 may adjust the magnitude and durationof partial-discharge pulses to achieve minimum power loss in the powerconverter and to guarantee that output rectifier D1 114 conducts inaccordance with the teachings of the present invention. This adjustmentmay be done, for example, in response to an external signal received bycontroller 144. The adjustment may also be done by choosing values ofdiscrete components within control circuit 144. In examples wherecontrol circuit 144 is included in an integrated circuit, the adjustmentmay be done by trimming the internal parameters of the integratedcircuit.

FIG. 4 is a schematic diagram of a power converter 400 that illustratesone example of current controller 148 in greater detail. In the exampleof FIG. 4, current controller 148 includes a mode selector 410 thatreceives mode select signal 146 from controller 142. In the example ofFIG. 4, mode selector 410 closes either switch Si 420, or switch S2 440,or neither switch in response to mode select signal 146.

In the example of FIG. 4, the first mode opens switch S1 420 and switchS2 440 such that current I_(P) 126 is substantially zero when currentcontroller 148 is in the first mode. In the example of FIG. 4, thesecond mode closes switch S1 420 and opens switch S2 440 to fullydischarge capacitor C_(P) 150. In the example of FIG. 4, the third modeopens switch S1 420 and closes switch S2 440 such that current I_(P) 126is the value of current source 430 to partially discharge capacitorC_(P) 150 in accordance with the teachings of the present invention. Itis appreciated that in other examples current source 430 could be avariable current source that varies in accordance with the degree ofpartial discharge of capacitor C_(P) 150 desired.

FIG. 5 is a schematic diagram of a power converter 500 that showsanother example of current controller 148 that includes a metal oxidesemiconductor field-effect transistor (MOSFET) 520 and a tri-leveldriver 510 to produce full-discharge and partial-discharge pulses ofcurrent I_(P) 126 in response to the mode select signal 146. In theexample of FIG. 5, tri-level driver 510 responds to signals from modeselector 410 to produce at least three distinct values of a voltagebetween the gate terminal and the source terminal of n-channel MOSFET520 in response to mode select signal 146. In one example, the gateterminal of MOSFET 520 may be considered as a control terminal of MOSFET520.

In the example of FIG. 5, the first mode of current controller 148applies a gate-to-source voltage substantially less than the thresholdvoltage of n-channel MOSFET 520. As a result, MOSFET 520 issubstantially switched OFF in the first mode of operation of currentcontroller 148. In the example of FIG. 5, the second mode of currentcontroller 148 applies a gate-to-source voltage substantially greaterthan the threshold voltage of n-channel MOSFET 520. As a result, MOSFET520 is substantially switched ON in the second mode of operation ofcurrent controller 148. In the example of FIG. 5, the third mode ofcurrent controller 148 applies a gate-to-source-voltage slightly higherthan the threshold voltage of n-channel MOSFET 520. As a result, thegate-to-source voltage at the control terminal of MOSFET 520 in thethird mode corresponds to MOSFET 520 providing a partial-dischargecurrent pulse for the magnitude of current I_(P) 126. In other words,when in the third mode of operation, MOSFET 520 operates not as a switchthat may be either open or closed, but in its saturation region,sometimes called the active region, where the drain current iscontrolled primarily by the gate-to-source voltage and is substantiallyindependent of the drain-to-source voltage. In examples where MOSFET 520and tri-level driver 510 are included in an integrated circuit,tri-level driver 510 may be designed such that the voltage applied tothe gate during the third mode of current controller 148 tracks thethreshold voltage of MOSFET 520, thereby reducing the change inpartial-discharge current over a range of temperature and processvariations. It is appreciated that in other examples, tri-level driver510 could have four or more drive levels to select different MOSFET 520saturation characteristics according to the degree of partial dischargeof capacitor C_(P) 150 desired.

FIG. 6 is a flow diagram 600 that shows one example method to control apower converter in accordance with the teachings of the presentinvention providing for sensing of an isolated output. After starting inblock 605, a current controller is operated in full capacitancedischarge mode in block 610 to produce a current pulse that fullydischarges capacitance on a node of the current controller.

Next, a first feedback circuit senses the isolated output voltage inblock 615. In block 620, information from feedback circuits is processedto estimate the condition of the load. Then, in decision block 625, theflow continues to block 630 if the load is near zero, or branches backto block 610 if the load is not near zero. In one example, the load isconsidered near zero when the full-discharge pulses occur at intervalsgreater than the wake-up period T_(W1) of FIG. 3.

In block 630, the current controller is operated in partial capacitancedischarge mode to produce a current pulse that only partially dischargesa capacitance on a node of the current controller. Then a secondfeedback circuit senses the isolated output voltage in block 635 beforereturning to block 620 where the information from feedback circuits isprocessed.

The above description of illustrated examples of the present invention,including what is described in the Abstract, are not intended to beexhaustive or to be limitation to the precise forms disclosed. Whilespecific embodiments of, and examples for, the invention are describedherein for illustrative purposes, various equivalent modifications arepossible without departing from the broader spirit and scope of thepresent invention. Indeed, it is appreciated that the specific voltages,currents, frequencies, power range values, times, etc., are provided forexplanation purposes and that other values may also be employed in otherembodiments and examples in accordance with the teachings of the presentinvention.

These modifications can be made to examples of the invention in light ofthe above detailed description. The terms used in the following claimsshould not be construed to limit the invention to the specificembodiments disclosed in the specification and the claims. Rather, thescope is to be determined entirely by the following claims, which are tobe construed in accordance with established doctrines of claiminterpretation. The present specification and figures are accordingly tobe regarded as illustrative rather than restrictive.

What is claimed is:
 1. A power converter, comprising: an energy transferelement; a current controller coupled to the energy transfer element andcoupled to an input of the power converter, wherein the currentcontroller is coupled to selectively enable one of a first current,second current or third current in the current controller, wherein thefirst current is substantially zero, the second current is greater thanthe third current, and the third current is greater than the firstcurrent, wherein the third current is controlled to only partiallydischarge a capacitance coupled to a terminal coupled between the energytransfer element and the current controller; a control circuit to becoupled to the current controller to selectively enable said one of thefirst current, second current or third current in the currentcontroller; a first feedback circuit coupled to the control circuit andcoupled to generate a first feedback signal representative of an outputof the power converter while the first current is enabled by the currentcontroller after a full discharge pulse of current in the currentcontroller; and a second feedback circuit coupled to the control circuitand coupled to generate a second feedback signal representative of theoutput of the power converter while the first current is enabled in thecontroller after a partial discharge pulse of current in the currentcontroller.
 2. The power converter of claim 1 wherein the controlcircuit is coupled to enable the first, second or third currents in thecurrent controller in response to the first and second feedback signalsto control a transfer of energy from the input of the power converter tothe output of the power converter.
 3. The power converter of claim 1wherein the current controller comprises a transistor coupled to theenergy transfer element and an input of the power converter.
 4. Thepower converter of claim 1 wherein the current controller comprises atri-level driver to provide a voltage having one of first, second orthird values in response to the control circuit to a control terminal ofa transistor coupled to the energy transfer element and an input of thepower converter.
 5. The power converter of claim 1 wherein the currentcontroller comprises a transistor coupled to be OFF when the firstcurrent is enabled in the current controller.
 6. The power converter ofclaim 1 wherein the current controller comprises a transistor coupled tobe ON when the second current is enabled in the current controller. 7.The power converter of claim 1 wherein the current controller comprisesa MOSFET coupled to operate in a saturation region of the MOSFET whenthe third current is enabled in the current controller.
 8. The powerconverter of claim 1 wherein the first feedback circuit is coupled tothe energy transfer element to generate the first feedback signal inresponse to a reflected signal representative of the output of the powerconverter.
 9. The power converter of claim 1 wherein the second feedbackcircuit is coupled to the energy transfer element to generate the secondfeedback signal in response to a portion of a decaying oscillation in areflected signal representative of the output of the power converter.10. The power converter of claim 1 wherein the full discharge pulse ofcurrent in the current controller occurs in response to the controlcircuit enabling the second current in the current controller followedby the control circuit enabling the first current in the currentcontroller.
 11. The power converter of claim 1 wherein the partialdischarge pulse of current in the current controller occurs in responseto the control circuit enabling the third current in the currentcontroller followed by the control circuit enabling the first current inthe current controller.